Method of controlling power conversion apparatus

ABSTRACT

A voltage of a reactor taking a potential on a side of a capacitor as a reference is detected. A correction coefficient which is made smaller as a first voltage control rate command becomes smaller is calculated, the first voltage control rate being a ratio of an amplitude of an AC voltage, which is outputted by a power converter, to an average value of the DC voltage. A correction is made to subtract a correction amount obtained by multiplying the voltage of the reactor by the correction coefficient from the first voltage control rate command, so that a second voltage control rate command is generated. A switching signal which is generated based on a second voltage control rate command is given to the power converter.

TECHNICAL FIELD

The present disclosure relates to a method of controlling a powerconversion apparatus, and for example, a controller for a capacitor-lessinverter.

BACKGROUND ART

Japanese Patent No. 4067021 describes an electric motor controller. Theelectric motor controller has a converter and an inverter. The converterand the inverter are connected to each other through a DC link. Theconverter inputs thereto an AC (alternating current) voltage, performs afull-wave rectification thereon for conversion, and outputs a DC (directcurrent) voltage to the DC link. The inverter inputs thereto the DCvoltage and performs conversion to output an AC voltage to an electricmotor.

Further, the DC link is provided with an LC filter having a reactor anda capacitor. In more detail, the capacitor and the reactor are connectedin series to each other between a pair of output ends of the converter,and a voltage across both ends of the capacitor is inputted to theinverter as the DC voltage. A capacitance of the capacitor is smallerthan that of a so-called smoothing capacitor, and the voltage acrossboth ends of the capacitor has a pulsating component caused by thefull-wave rectification.

In Japanese Patent No. 4067021, in order to reduce a harmonic componentof DC voltage caused by resonance of the LC filter, the inverter iscontrolled on the basis of a voltage across both ends of the reactor.For example, a correction of subtracting a value, obtained bymultiplying the voltage of the reactor by a gain, from an initial valuefor a voltage control rate of the inverter is made to calculate a targetvalue of a voltage control rate command. Then, a switching signal to theinverter is generated on the basis of the target value of the voltagecontrol rate and a voltage command value calculated by a known method.This reduces the harmonic component of the voltage across both ends ofthe capacitor and further reduces distortion of the current to beinputted to the electric motor controller. The control based on areactor voltage is referred to also as a “VL control system” in thepresent application.

SUMMARY OF INVENTION Problems to be Solved by the Invention

When the initial value of the voltage control rate is small, the gain isrelatively large. That is to say, a ratio of the gain to the initialvalue of the voltage control rate increases. Too large ratio may causean instability of a VL control system.

Then, it is an object of the present disclosure to provide a method ofcontrolling a power conversion apparatus, which is capable ofsuppressing an instability of a VL control system caused by an increasein a ratio of a gain to a voltage control rate.

Means for Solving the Problems

The present disclosure is intended for a method of controlling a powerconversion apparatus which includes a first power supply line (LH) and asecond power supply line (LL), a rectifier (1) for rectifying aninputted first AC voltage into a DC voltage to output the DC voltage tobetween the first power supply line and the second power supply line, acapacitor (C1) provided between the first power supply line and thesecond power supply line, a reactor (L1) constituting an LC filter,together with the capacitor, and a power converter (2) for converting aDC voltage (Vdc) supported by the capacitor into a second AC voltagebased on a switching signal (S) to be inputted and then applying thesecond AC voltage to a load to flow an alternating current. According toa first aspect of the present disclosure, the method of controlling thepower conversion apparatus includes steps of detecting a voltage (VL) ofthe reactor taking a potential on a side of the capacitor as areference; when at least a first voltage control rate command (ks**)which is a ratio of an amplitude of the second AC voltage to an averagevalue of the DC voltage is smaller than a predetermined value,calculating a correction coefficient which is made smaller as the firstvoltage control rate command becomes smaller;

making a correction on the first voltage control rate command tosubtract a correction amount (H) obtained by multiplying the voltage ofthe reactor by the correction coefficient, thereby generating a secondvoltage control rate command (ks**); and giving the switching signalwhich is generated based on the second voltage control rate command tothe power converter.

According to a second aspect of the present disclosure, in the method ofcontrolling the power conversion apparatus of the first aspect, when thefirst voltage control rate command is larger than the predeterminedvalue, the correction coefficient is calculated to be made larger as theamplitude of the alternating current becomes smaller.

According to a third aspect of the present disclosure, in the method ofcontrolling the power conversion apparatus of the second aspect, whenthe first voltage control rate command is larger than the predeterminedvalue, the correction coefficient is inversely proportional to theamplitude of the alternating current.

According to a fourth aspect of the present disclosure, in the method ofcontrolling the power conversion apparatus of any one of the first tothird aspects, when the correction amount is larger than a positiveupper limit value, the correction amount is limited to the upper limitvalue, and when said correction amount is smaller than a negative lowerlimit value, the correction amount is limited to the lower limit value,and absolute values of the upper limit value and lower limit value aremade larger as the first voltage control rate command becomes larger.

According to a fifth aspect of the present disclosure, in the method ofcontrolling the power conversion apparatus of the fourth aspect, theupper limit value or the lower limit value is proportional to the firstvoltage control rate command.

Effects of the Invention

In the method of controlling a power conversion apparatus according tothe first aspect of the present disclosure, the correction amountobtained by multiplying the voltage of the reactor by the correctioncoefficient is subtracted from the first voltage control rate command,so that this correction coefficient functions as a gain when the reactorvoltage is fed back.

The first voltage control rate command is made small as the amplitude ofthe alternating current becomes small, and the correction coefficient isalso made small as the amplitude of the alternating current becomessmall.

Accordingly, an increase in a ratio of the correction coefficient (gain)to the first voltage control rate command can be suppressed, thus theinstability of the control can be suppressed.

In the method of controlling a power conversion apparatus according tothe second aspect of the present disclosure, when the amplitude of thealternating current is large, the first voltage control rate command isalso set to be large, so that when the amplitude of the alternatingcurrent is larger than the predetermined value, the ratio of the gain tothe first voltage control rate command is less large compared with thecase where the alternating current is smaller than the predeterminedvalue. Accordingly, the instability of the control due to the excessivegain does not easily occur.

In the meanwhile, when the amplitude of the alternating current islarger than the predetermined value, a transfer gain of a VL controlsystem increases in accordance with the increase in the amplitude of thealternating current (described in detail in the embodiment), andaccordingly, a gain margin decreases and the instability of the controlmay occur.

According to the second aspect of the present disclosure, when theamplitude of the alternating current is larger than the predeterminedvalue, the correction coefficient is calculated to be made larger as theamplitude of the alternating current becomes smaller. Accordingly, avariation of the gain of the VL control system due to a variation of theamplitude of the alternating current can be suppressed, so thatinstability of the load can be suppressed (described in detail in theembodiment).

In the method of controlling a power conversion apparatus according tothe third aspect of the present disclosure, the variation of the gaindue to the variation of the amplitude of the alternating current can betheoretically resolved.

In the method of controlling a power conversion apparatus according tothe fourth aspect of the present disclosure, the ratio of the correctionamount to the first voltage control command can be prevented frombecoming too large (exceeding the upper limit value).

In the method of controlling a power conversion apparatus according tothe fifth aspect of the present disclosure, the upper limit value or thelower limit value can be easily generated.

These and other objects, features, aspects and advantages of the presentdisclosure will become more apparent from the following detaileddescription of the present disclosure when taken in conjunction with theaccompanying drawings.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a view showing an exemplary conceptual configuration of apower conversion apparatus;

FIG. 2 is a view showing an exemplary conceptual configuration of acontroller;

FIG. 3 is a view showing an exemplary equivalent circuit;

FIG. 4 is a view showing an exemplary block diagram;

FIG. 5 is a view showing an exemplary block diagram;

FIG. 6 is a view showing an exemplary block diagram;

FIG. 7 is a view showing an exemplary equivalent circuit;

FIG. 8 is a view showing an exemplary block diagram;

FIG. 9 is a view showing an exemplary block diagram;

FIG. 10 is a view showing exemplary transfer gain and phase of anopen-loop transfer function;

FIG. 11 is a view showing an exemplary block diagram;

FIG. 12 is a view showing an exemplary block diagram;

FIG. 13 is a view showing an exemplary block diagram;

FIG. 14 is a view showing an exemplary relationship between a correctioncoefficient and an amplitude of an alternating current;

FIG. 15 is a view showing exemplary input current and direct current;

FIG. 16 is a view showing exemplary input current and direct current;

FIG. 17 is a view showing an exemplary conceptual configuration of acontroller; and

FIG. 18 is a view showing an exemplary relationship between a limitvalue and a voltage control rate command for a correction amount.

DESCRIPTION OF EMBODIMENT(S) The First Embodiment

<1. Configuration of Power Conversion Apparatus>

As shown in FIG. 1, a present power conversion apparatus includes arectifier 1, a capacitor C1, a reactor L1, and a power converter 2.

The rectifier 1 converts an N-phase AC (alternating current) voltage (N:natural number) inputted from an AC (alternating current) power supplyE1 into a DC (direct current) voltage and outputs the DC voltage tobetween DC (direct current) lines (power supply lines) LH and LL. In theexemplary case of FIG. 1, the rectifier 1 is a diode rectifier circuit.The rectifier 1 is not limited to the diode rectifier circuit but may bea separately-excited rectifier circuit or a self-excited rectifiercircuit. As the separately-excited rectifier circuit, for example, athyristor bridge rectifier circuit can be adopted, and as theself-excited rectifier circuit, a PWM (Pulse-Width-Modulation) AC-DCconverter can be adopted.

Further, in the exemplary case of FIG. 1, the rectifier 1 is athree-phase rectifier circuit to which a three-phase AC voltage isinputted. The number of phases of the AC voltage to be inputted to therectifier 1, i.e., the number of phases of the rectifier 1, however, isnot limited to three but may be set as appropriate.

The capacitor C1 is provided between the DC lines LH and LL. Thecapacitor C1 is, for example, a film capacitor. Such a capacitor C1 ischeaper than an electrolytic capacitor. On the other hand, thecapacitance of the capacitor C1 is smaller than that of the electrolyticcapacitor, so that the capacitor C1 cannot sufficiently smooth the DCvoltage Vdc between the DC lines LH and LL. In other words, thecapacitor C1 allows the pulsation of a rectified voltage which isrectified by the rectifier 1. Therefore, the DC voltage Vdc has apulsating component caused by the rectification of the N-phase ACvoltage (a pulsating component having a frequency which is 2N times ashigh as that of the N-phase AC voltage in case of using the full-waverectification, for example). In the exemplary case of FIG. 1, since thethree-phase AC voltage is full-wave rectified, the DC voltage Vdcpulsates at a frequency which is six times as high as that of thethree-phase AC voltage.

The reactor L1 constitutes an LC filter, together with the capacitor C1.In the exemplary case of FIG. 1, the reactor L1 is provided in the DCline LH or LL (in the exemplary case of FIG. 1, in the DC line LH) onthe side of the rectifier 1 from the capacitor C1. The location of thereactor L1 is not limited to this, but the reactor L1 may be provided onthe side of the input of the rectifier 1.

The reactor L1 and the capacitor C1, which are connected in series toeach other between a pair of output ends of the AC power supply E1,constitute a so-called LC filter. When the capacitance of the capacitorC1 is small as described above, a resonance frequency of this LC filtertends to be higher. Similarly, as the inductance of the reactor L1 ismade smaller, the resonance frequency tends to become still higher. In acase, for example, where the capacitance of the capacitor C1 is 40 μFand the inductance of the reactor L1 is 0.5 mH in FIG. 1, the resonancefrequency is about 1.125 kHz.

The power converter 2 is, for example, a voltage type inverter andinputs thereto the DC voltage Vdc between the DC lines LH and LL (the DCvoltage supported by the capacitor C1). Then, the power converter 2converts the DC voltage Vdc into an AC voltage on the basis of aswitching signal S from a controller 3 and outputs the AC voltage to aload M1. Hereinafter, the AC voltage outputted from the power converter2 is also referred to as an output voltage.

In FIG. 1, for example, the power converter 2 has three pairs ofswitching units (for three phases), each pair of which are connected inseries to each other between the DC lines LH and LL. In the exemplarycase of FIG. 1, a pair of switching units Sup and Sun are connected inseries to each other, another pair of switching units Svp and Svn areconnected in series to each other, and still another pair of switchingunits Swp and Swn are connected in series to each other. Then, a nodebetween a pair of switching units Sxp and Sxn (x represents any one ofu, v, and w, the same applies to the following) for each phase isconnected to the load M1 through an output line Px. When these switchingunits Sxp and Sxn become conducting/nonconducting on the basis of anappropriate switching signal, the power converter 2 converts the DCvoltage Vdc into a three-phase AC voltage and outputs the AC voltage tothe load M1. Accordingly, the alternating current flows through the loadM1.

The load M1 may be, for example, a rotary machine (e.g., an inductionmachine or a synchronous machine). Though the three-phase load M1 isshown in the exemplary case of FIG. 1, the number of phases is notlimited to three. In other words, the power converter 2 is not limitedto a three-phase power converter.

<2. Control>

Herein, the control over the power converter 2 is performed byintroducing a voltage control rate ks. The above voltage control rate ksrefers to a ratio of an amplitude Vm of the output voltage from thepower converter 2 to the DC voltage Vdc (=Vm/Vdc). In other words, thevoltage control rate ks indicates a ratio at which the AC voltage isoutputted with respect to the DC voltage Vdc.

Since the power converter 2 performs a switching operation, the DCvoltage Vdc varies with the switching. In other words, a harmoniccomponent is generated in the DC voltage Vdc. Further, since a switchingfrequency is higher than the frequency of the pulsation of the DCvoltage Vdc (hereinafter, referred to also as a pulsation frequency)caused by the rectification, the frequency of the harmonic componentwhich is described herein is higher than the pulsation frequency.Furthermore, the present power conversion apparatus has the LC filterconstituted of the capacitor C1 and the reactor L1 as described above.Therefore, as the switching frequency becomes closer to the resonancefrequency of the LC filter, the range of variation in the harmoniccomponent of the DC voltage Vdc in the capacitor C1 increases.

Such a harmonic component of the DC voltage Vdc is not desirable becausethe harmonic component invites, for example, a harmonic component of analternating current to be inputted to the rectifier 1.

Then, in order to reduce the harmonic component of the DC voltage Vdc,the voltage control rate ks is corrected. In more detail, a correctionis made so that the voltage control rate ks can increase as the harmoniccomponent of the DC voltage Vdc becomes larger. With this correction,the amplitude Vm of the output voltage increases as the harmoniccomponent of the DC voltage Vdc becomes higher. Thereby, since theoutput power of the power converter 2 becomes higher, the DC voltage Vdcdecreases. Therefore, the harmonic component of the DC voltage Vdc canbe reduced.

Such a correction is made on the basis of a voltage VL supported by thereactor L1. The reason is that when the N-phase AC voltage to beinputted to the rectifier 1 is assumed to be an ideal voltage source, itcan be considered that the harmonic component of the DC voltage Vdcappears as the voltage VL. The harmonic component which appears in thevoltage VL, however, is in phase or in opposite phase with the harmoniccomponent of the DC voltage Vdc, depending on how to take a referencepotential of the voltage VL. When the potential on the side of thecapacitor C1 of the reactor L1 in FIG. 1 is taken as a reference, forexample, a harmonic component in opposite phase with the harmoniccomponent of the DC voltage Vdc appears in the voltage VL. When thepotential on the opposite side of the capacitor C1 (in other words, onthe side of the rectifier 1) of the reactor L1 is taken as a reference,a harmonic component in phase with the harmonic component of the DCvoltage Vdc appears in the voltage VL.

Therefore, when the potential on the side of the capacitor C1 is takenas a reference, for example, a correction is made so that the voltagecontrol rate ks can decrease as the voltage VL becomes higher, and whenthe potential on the opposite side is taken as a reference, a correctionis made so that the voltage control rate ks can increase as the voltageVL becomes higher. The correction can be thereby made so that thevoltage control rate ks can increase as the harmonic component of the DCvoltage Vdc becomes higher.

A conventional correction of the voltage control rate ks is describedmore specifically in case of adopting the voltage VL taking thepotential on the side of the capacitor C1 as a reference. A correctedvoltage control rate command ks* is calculated by subtracting acorrection amount H obtained by multiplying the voltage VL by apredetermined gain K (=K·VL) from an uncorrected voltage control ratecommand ks**. When this is formulated, the following equation isderived.ks*=ks**−K·VL  (1)

Accordingly, the voltage control rate command ks* can decrease as thevoltage VL becomes higher. Therefore, a correction can be made so thatthe voltage control rate command ks* is made larger as the harmoniccomponent of the DC voltage Vdc becomes larger.

The gain K is a parameter to determine a ratio at which the voltagecontrol rate command ks* decreases with respect to the value of thevoltage VL, in other words, the harmonic component of the DC voltageVdc. The correction amount (a reduction amount of the voltage controlrate command ks*) is made larger as the gain K becomes larger.

When the voltage VL taking the potential on the side of the rectifier 1as a reference is adopted, the corrected voltage control rate commandks* is calculated by adding the correction amount H to the uncorrectedvoltage control rate command ks**. When this is formulated, thefollowing equation is derived.ks*=ks**+K·VL  (2)

The ratio of the gain K to the voltage control rate command ks**(K/ks**) is made larger as the voltage control rate command ks** becomessmaller. When the gain K is too large for the voltage control ratecommand ks**, a stability of the VL control system may be impaired.

Thus, in the present embodiment, a correction coefficient α which ismade smaller as the voltage control rate command ks** becomes smaller isintroduced. In more detail, a product of the voltage VL, the gain K, andthe correction coefficient α is taken as the correction amount H, andthis correction amount H is subtracted from the voltage control ratecommand ks**. When this is formulated, the following equation isderived. Although a case where the potential on the side of thecapacitor C1 is taken as the reference of the voltage VL of the reactoris described below as an exemplary case, the equation (2) may also beused.ks*=ks**−α·K·VL  (3)

The product of the correction coefficient α and the gain K, which servesas the coefficient of the voltage VL, may be understood as thecorrection coefficient (α·K). This correction coefficient (α·K) alsoserves as a gain of the voltage VL. Herein, the correction coefficient(α·K) is also made smaller as the voltage control rate command ks**becomes smaller.

According to the equation (3), the correction coefficient (α·K) is alsomade smaller as the voltage control rate command ks** becomes smaller.Thus, an increase in the ratio of the correction coefficient to thevoltage control rate command ks** (=α·K/ks**) can be suppressed comparedwith the conventional technique. In other words, the correctioncoefficient can be suppressed from becoming too large for the voltagecontrol rate command ks**. Thus, the instability of the control can besuppressed.

A value proportional to the voltage control rate command ks**, forexample, may be taken as the correction coefficient α. According to thiscase, the correction coefficient α can be simply calculated.

<3. Control Configuration>

A specific control configuration is described. As shown in FIG. 1, thepresent power conversion apparatus is provided with a reactor voltagedetector 4. The reactor voltage detector 4 detects the voltage VL of thereactor L1, and performs, for example, an analog-digital conversion onthe voltage VL to output the converted voltage VL to the controller 3.Herein, as an exemplary case, the voltage VL is a voltage taking apotential on the side of the capacitor C1 among the potentials of bothends of the reactor L1 as a reference. The voltage VL detected by thereactor voltage detector 4 is used for the correction of the voltagecontrol rate ks.

The present power conversion apparatus is also provided with a currentdetector 5. The current detector 5 detects the alternating currentoutputted by the power converter 2 (the alternating current flowingthrough the load M1), and performs, for example, an analog-digitalconversion on the alternating current to output the convertedalternating current to the controller 3. The exemplary case of FIG. 1shows that the power converter 2 outputs three phase (U-phase, V-phase,and W-phase) alternating currents and the two phase (U-phase andV-phase) alternating currents iu and iv are detected among three phasealternating current. Since a sum of the three phase alternating currentsis ideally to be zero, the controller 3 can calculate the remaining onephase alternating current iw from the two phase alternating currents iuand iv. These currents are appropriately used by a known method togenerate the switching signal S.

As shown in FIG. 1, the controller 3 includes a harmonic suppressioncontroller 31, a voltage control rate corrector 32, and a switchingsignal generator 33.

The controller 3 includes, for example, a microcomputer and a memorydevice. The microcomputer executes process steps (in other words,procedures) described in a program. The above memory device can beconstituted of one or a plurality of memory devices such as a ROM (ReadOnly Memory), a RAM (Random Access Memory), a rewritable nonvolatilememory (EPROM (Erasable Programmable ROM) or the like), a hard diskunit, and the like. The memory device stores therein various informationand data and the like, also stores therein a program to be executed bythe microcomputer, and provides a work area for execution of theprogram. Further, it can be grasped that the microcomputer functions asvarious means corresponding to the process steps described in theprogram, or that the microcomputer implements various functionscorresponding to the process steps. Furthermore, the controller 3 is notlimited to this type but some or all of the procedures executed by thecontroller 3, or some or all of the means or functions implemented bythe controller 3 may be achieved by hardware.

The harmonic suppression controller 31 is a function unit to suppressthe harmonic component of the DC voltage Vdc. The harmonic suppressioncontroller 31 receives the voltage control rate command ks** and thevoltage VL and calculates, on the basis of them, the above correctionamount H used for the harmonic suppression.

FIG. 2 is a functional block diagram showing an exemplary case of aparticular internal configuration of the harmonic suppression controller31. The harmonic suppression controller 31 includes a correctioncoefficient calculator 311, a gain unit 312, and a multiplier 313.

The voltage control rate command ks** is inputted to the correctioncoefficient calculator 311. The correction coefficient α calculated bythe correction coefficient calculator 311 is made smaller as the voltagecontrol rate command ks** becomes smaller. For example, the voltagecontrol rate command ks** is multiplied by a predeterminedproportionality coefficient (>0) so that the correction coefficient α iscalculated. This correction coefficient α is outputted to the multiplier313.

The gain unit 312 receives the voltage VL. Then, the gain unit 312multiplies the voltage VL by the gain K and output the result to themultiplier 313.

The multiplier 313 receives the result (K·VL) outputted from the gainunit 312 and the correction coefficient α. Then the multiplier 313multiplies the result by the correction coefficient α to calculate thecorrection amount H (=α·K·VL) and then outputs this correction amount Hto the voltage control rate corrector 32.

With reference to FIG. 1 again, the voltage control rate corrector 32inputs the correction amount H and the voltage control rate command ks**and outputs the voltage control rate command ks*. The voltage controlrate corrector 32 subtracts the correction amount H from the voltagecontrol rate command ks** to calculate the voltage control rate commandks*. The voltage control rate command ks* is outputted to the switchingsignal generator 33.

The switching signal generator 33 generates, on the basis of awell-known method (for example, Japanese Patent No. 4067021), a voltagecommand (for example, a d-axis voltage command and a q-axis voltagecommand in a d-q axis rotating coordinate system) for the AC voltageoutputted by the power converter 2. Then, for example, a correction ismade by multiplying the voltage command by the voltage control ratecommand ks*. Subsequently, the switching signal S is generated by awell-known method (for example, a method of performing a coordinateconversion on the corrected voltage command so that the correctedvoltage command is converted into the three-phase voltage command, andthen comparing the three-phase voltage command and a carrier) based onthe corrected voltage command. The switching signal S is outputted tothe power converter 2.

Accordingly, the AC voltage based on the voltage control rate commandks* is outputted. Thus, the harmonic component of the DC voltage Vdc canbe suppressed. Moreover, since the correction coefficient α (or α·K) ismade smaller as the voltage control rate command ks** becomes smaller,an increase in the ratio of the correction coefficient α (or α·K) to thevoltage control rate command ks** can be suppressed. Thus, theinstability of the control caused by the increase can be suppressed.

The Second Embodiment

The second embodiment is different from the first embodiment in thecalculation method of the correction coefficient α. In the secondembodiment, when the voltage control rate command ks** is smaller than apredetermined value, the correction coefficient α is calculated to bemade smaller as the voltage control rate command ks** becomes smaller asis the case with the first embodiment, and in the meanwhile, when thevoltage control rate command ks** is larger than the predeterminedvalue, the correction coefficient α is calculated to be made larger asthe amplitude of the alternating current becomes smaller. Thepredetermined value is previously determined by, for example, anexperiment or a simulation.

In a region where the voltage control rate command ks** is larger thanthe predetermined value (hereinafter, referred to also as a high loadregion), the ratio of the correction coefficient (the gain K) to thevoltage control rate command ks** is comparatively small even when theconventional equation (1) is adopted. Thus, in the high load region, theincrease of the ratio does not impair the stability of the VL controlsystem easily.

In the meanwhile, the stability of the VL control system in the highload region is largely influenced by the variation of gain (the gainhere indicates a largeness of a transfer function, referred to atransfer gain hereinafter) in the VL control system in accordance withthe variation of amplitude of the alternating current. The transfer gainis described in detail hereinafter.

Hereinafter, a case of α=1 is described first, and then a case of α=α2(α2 is made larger as the amplitude of the alternating current becomessmaller) is described, for the explanation in an orderly sequence.

FIG. 3 shows a simplified equivalent circuit in the power conversionapparatus of FIG. 1. Herein, the transfer function in an equivalentcircuit of FIG. 7 is derived in consideration of the transfer functionusing the simplified equivalent circuit of FIG. 3. In the exemplary caseof FIG. 3, it is grasped that the load M1 is an inductive load and asubsequent stage after the power converter 2 is a current source.Further, since the source impedance between the AC power supply E1 andthe rectifier 1 also has an effect on the resonance frequency of the LCfilter, the source impedance is also shown in the equivalent circuit ofFIG. 3.

Herein, the resistance value and the inductance of the source impedance,the inductance of the reactor L1, and the capacitance of the capacitorC1 are represented by “r”, “l”, “L”, and “C”, respectively. A currentflowing in the reactor L1, a current flowing in the capacitor C1, and acurrent flowing in the current source are represented by “IL”, “Ic”, and“Io”, respectively. These quantities are indicated near thecorresponding constituent elements in FIG. 3.

FIG. 4 shows a block diagram of the VL control system. In the presentcontrol method, the harmonic component (in opposite phase with thevoltage VL) of the DC voltage Vdc is reduced by the correction of thevoltage control rate command on the basis of the voltage VL. Therefore,there is a concept of feedback in which a control is performed toapproximate the voltage VL to a target value. Herein, since thestability of the VL control system is taken into consideration, FIG. 4shows a block diagram of a feedback control system for the voltage VL.

An exemplary case of FIG. 4 shows “K·e^(−st)” as a dead time element.This exemplary case shows an element caused by a dead time t from a timewhen the voltage VL is detected until the control based on the voltageVL is reflected. In other words, the dead time t shows a time periodfrom the time when the voltage VL is detected until the power converter2 outputs the AC voltage based on the switching signal S which isgenerated using the voltage control rate command ks* (=ks**−K·VL).

When a well-known conversion is performed on the block diagram of FIG.4, the block diagram of FIG. 5 is derived. Thus, an open-loop transferfunction G0 is shown by a block diagram of FIG. 6. The open-looptransfer function G0 is a product of equations indicated by the twoelements in FIG. 6.

Next, as shown in FIG. 7, it is grasped that the power converter 2 isdivided into a current source and a voltage source. The load M1, whichis the inductive load, is grasped as a current source.

In the equivalent circuit, the DC voltage Vdc to be inputted to thepower converter 2 and the amplitude Vm of the output voltage of thepower converter 2 satisfy the following equation.Vm=(ks−K·VL)·Vdc  (4)

Further, ideally, an electric power on the input side of the powerconverter 2 and an electric power on the output side thereof are equalto each other. Herein, when a power factor on the output side of thepower converter 2 (a so-called load power factor) is assumed to be 1,for simplification, the following equation is true.√{square root over ( )}3·Vrms·Irms=Vdc·Idc  (5)

In the equation (5), “Vrms” represents an effective value of the outputvoltage of the power converter 2, “Irms” represents an effective valueof the output current (the alternating current flowing through the loadM1) of the power converter 2, and “Idc” represents the direct current tobe inputted to the power converter 2. Herein, as an exemplary case, itis assumed that the power converter 2 outputs a three-phase AC voltage.Therefore, in the left side of the equation (5), √{square root over ()}3 is present as a factor. Further, a current Io1 of the equivalentcircuit is grasped as an effective value and understood to be equal tothe effective value Irms. In the block diagram described later, thecurrent Io1 of the equivalent circuit is used instead of the effectivevalue Ims.

Further, the amplitude Vm and the effective value Vrms of the outputvoltage satisfy the following equation.Vm=√{square root over ( )}2·Vrms  (6)

When the effective value Vrms and the amplitude Vm are deleted by usingthe equations (4) to (6), the following equation is derived.√{square root over ( )}(3/2)·(ks−K·VL)·Vdc·Irms=Vdc·Idc  (7)

When both sides are multiplied by a reciprocal of the DC voltage Vdc,respectively, the following equation is derived.√{square root over ( )}(3/2)·ks·Irms−√{square root over ()}(3/2)·Irms·(K·VL)=Idc  (8)

The first term in the left side of the equation (8) is a certaincomponent of the direct current Idc and is the direct current Idc in acase where the correction is not made on the voltage control ratecommand. Therefore, when the direct current Idc in a case where thecorrection is made on the voltage control rate command is represented asa direct current Idc′ to distinguish it from one on which the correctionis not made, the following equation is derived.Idc−√{square root over ( )}(3/2)·Irms·(K·VL)=Idc′  (9)

Since the second term in the left side of the equation (9) includes acorrection amount (K·VL) for the voltage control rate ks as a factor,the second term is a variation component caused by the correction basedon the voltage VL. Further, since the harmonic component of the DCvoltage Vdc appears in the voltage VL, the second term can be grasped asa variation component based on the harmonic component of the DC voltageVdc. The second term also includes the effective value Irms of theoutput current as a factor.

Thus, the correction of the voltage control rate ks on the basis of thevoltage VL means the correction on the direct current Idc. Then, thedirect current Idc is necessarily affected by the effective value Irms.Specifically, the correction is made by subtracting a result (product)obtained by multiplying the correction value (K·VL) based on the voltageVL by the coefficient √{square root over ( )}(3/2)·Irms based on theeffective value Irms from the direct current Idc.

Since the value obtained by multiplying the value (K·VL) by √{squareroot over ( )}(3/2)·Irms is the correction amount for the direct currentIdc, the block diagram of the VL control system has a configuration inwhich the element of “√{square root over ( )}(3/2)·Io1” (herein,Io1=Irms) is added to the block diagram of FIG. 5, as shown in FIG. 8.When the block diagram of FIG. 8 is converted in order to obtain anopen-loop transfer function G0′, the block diagram of FIG. 9 is derived.The block diagram of FIG. 9 have a configuration in which an element of“√{square root over ( )}(3/2)·Io1” is added to the block diagrams ofFIG. 6. Hereinafter, transfer functions having elements of “K·e^(−st)”,“√{square root over ( )}(3/2)·Io1”, and “(L·s/C)/{(L+1) s²+r·s+1/C}” arereferred to as G1 to G3, respectively.

As can be understood from the block diagram of FIG. 9, a transfer gain(dB) of the open-loop transfer function G0′ is a sum of the transfergains (dB) of the transfer functions G1 to G3. Since the transferfunction G2 is proportional to the effective value Irms (=Io1), thetransfer gain of the open-loop transfer function G0′ varies with theeffective value Irms. FIG. 10 shows transfer gains and phases of theopen-loop transfer function G0′ when the effective value Irms is 5 A, 10A, and 20 A. In FIG. 10, the transfer gains when the effective valueIrms is 5 A, 10 A, and 20 A are indicated by a one-dot chain line, adotted line, and a solid line, respectively. Since the effective valueIrms is positive, the transfer gain increases as the effective valueIrms becomes larger, as shown in FIG. 10.

On the other hand, since the transfer function G2 is a real number, thephase thereof is 0 degree. Therefore, even when the effective value Irmsvaries, the phase of the open-loop transfer function G0′ is not affectedthereby. Thus, in FIG. 10, the phases of the open-loop transfer functionG0′ when the effective value Irms is 5 A, 10 A, and 20 A are indicatedby a solid line in common. Accordingly, the frequency f1 when the phaseof the open-loop transfer function G0′ takes −180 degrees does notdepend on the effective value Irms.

Thus, though the frequency f1 does not vary with the effective valueIrms, the transfer gain increases as the effective value Irms becomeslarger. Therefore, the gain margin (an absolute value of the transfergain when the phase takes −180 degrees) decreases as the effective valueIrms becomes larger, and this may invite instability of the control.

Then, in the second embodiment, the voltage control rate ks is correctedas follows in the high load region. Specifically, a correctioncoefficient α2 (<1) which increases as the effective value Irms of thealternating current becomes smaller is adopted. When this is expressedby an equation, the following equation is derived.ks*=ks**−α2·K·VL  (10)

By adopting such a correction, the equation (9) is changed to thefollowing equation. In other words, the equation (9) corresponds to thecase of adopting α=1 in the equation (3), and an equation (11)corresponds to the case of adopting α=α2 in the equation (3).Idc−√{square root over ( )}(3/2)·Irms·α2·(K·VL)=Idc′  (11)

Since a second term in the left side of the equation (12) is thecorrection amount, the block diagram of the VL control system in thecase where the above correction is adopted has a configuration in whichan element of “α2” is added to the block diagram of FIG. 8, as shown inFIG. 11. When the block diagram of FIG. 11 is converted in order toobtain the open-loop transfer function G0″, a block diagram of FIG. 12are derived. The block diagram of FIG. 12 has a configuration in whichthe element of “α2” is added to the block diagram of FIG. 9.Hereinafter, a transfer function of the above element will be referredto as a transfer function G4.

Therefore, the transfer gain of the open-loop transfer function G0″ is asum of the transfer gain of the transfer functions G1 to G4. Since thecorrection coefficient α2 is smaller than 1, the transfer gain of thetransfer function G4 has a negative value when a unit dB is adopted.Further, since the correction coefficient α2 takes a smaller value asthe effective value Irms becomes larger, the transfer gain of thetransfer function G4 also decreases as the effective value Irms becomeslarger. Therefore, even when the transfer gain of the transfer functionG2 increases as the increase of the effective value Irms, the transfergain of the transfer function G4 decreases, so that the increase of thetransfer gain of the open-loop transfer function G0″ can be suppressed.Accordingly, the reduction in the gain margin due to the increase in theeffective value Irms can be suppressed, and this can contribute to thestability of the control.

It is desirable that a reciprocal of the effective value Irms (1/Irms)should be adopted as the correction coefficient α2. In other words, itis desirable that a correction should be made by subtracting a productof the voltage VL, the gain K, and the reciprocal of the effective valueIrms (the correction amount H) from the voltage control rate commandks**. As can be understood from the block diagram of FIG. 12, it isthereby possible to cancel the effective value Irms by multiplication ofthe correction coefficient α2 and the effective value Irms. Therefore,in this case, it is possible to avoid the variation in the transfer gainof the open-loop transfer function G0″ due to the variation in theeffective value Irms. Accordingly, the gain margin does not decreaseeven when the effective value Irms increases, and this can contribute tothe stability of the control.

Further, even when the effective value Irms is cancelled as thecorrection coefficient α2 in the block diagram of FIG. 12, √{square rootover ( )}(3/2) remains. This can be regarded as an offset of thetransfer gain when the unit dB is adopted. In order to also cancel√{square root over ( )}(3/2), √{square root over ( )}(⅔)/Irms should beadopted as the correction coefficient α2. This makes the sum of thetransfer gains (dB) of the transfer functions G3 and G4 zero (the valueof the transfer gain is 1).

Further, √{square root over ( )}3 in the equation (12) arises from√{square root over ( )}3 in the equation (5). Therefore, when the powerconverter 2 outputs a single-phase AC voltage, √{square root over ()}2/Irms should be adopted as the correction coefficient α2.

As described above, in the high load region where the voltage controlrate command ks** is larger than the predetermined value, the correctioncoefficient α2 which is made larger as the amplitude of the alternatingcurrent becomes smaller is adopted. In other words, in the high loadregion where the ratio of the correction coefficient (α·K) to thevoltage control rate command ks** is comparatively small and thus theinstability due to the ratio does not easily occur, the correctioncoefficient α2 is adopted to suppress not the instability of the controldue to the ratio but the instability of the control due to the increasein the amplitude of the alternating current.

In the meanwhile, in the low load region where the voltage control ratecommand ks** is smaller than the predetermined value, the correctioncoefficient α described in the first embodiment is adopted to suppressthe instability of the control due to the ratio.

Accordingly, the second embodiment enables the stability of the controleffectively in the low load region and the high load region.

<Configuration>

The power conversion apparatus according to the second embodiment isdifferent from the power conversion apparatus of FIG. 1 in the internalconfiguration of the harmonic suppression controller. FIG. 13 is afunctional block diagram conceptually showing an exemplary case of aninternal configuration of the harmonic suppression controller 31according to the second embodiment. The harmonic suppression controller31 further includes a correction coefficient calculator 314 and adetermination unit 315 compared with the harmonic suppression controller31 of FIG. 2.

The correction coefficient calculator 314 receives the effective valueIrms of the alternating current flowing through the load M1. Thecorrection coefficient calculator 314 calculates the correctioncoefficient α2 which increases as the effective value Irms becomessmaller. For example, a correction coefficient α2 which is inverselyproportional to the effective value Irms is calculated.

Hereinafter, the correction coefficient α calculated by the correctioncoefficient calculator 311 is referred to as a correction coefficientα1, and the correction coefficient which sums up the correctioncoefficients at and α2 is referred to as the correction coefficient α.As described in the first embodiment, the correction coefficient α1 iscalculated to be made smaller as the voltage control rate command ks**becomes smaller. For example, the correction coefficient α1 isproportional to the voltage control rate command ks**.

The correction coefficients α1 and α2 are inputted to the determinationunit 315. The voltage control rate command ks** is also inputted to thedetermination unit 315. Then, the determination unit 315 selects one ofthe correction coefficients al and α2 in accordance with the voltagecontrol rate command ks**. In more detail, the determination unit 315selects the correction coefficient α1 when the voltage control ratecommand ks** is smaller than the predetermined value, and outputs thecorrection coefficient α1 as the correction coefficient α to themultiplier 313. In the meanwhile, the determination unit 315 selects thecorrection coefficient α2 when the voltage control rate command ks** islarger than the predetermined value, and outputs the correctioncoefficient α2 as the correction coefficient α to the multiplier 313.The predetermined value is previously determined by an experiment or asimulation and is stored in the determination unit 315, for example.

The multiplier 313 multiplies the result (K·VL) outputted from the gainunit 312 by the correction coefficient α in the same manner as the firstembodiment to calculate the correction amount H and then outputs thiscorrection amount H to the voltage control rate corrector 32.

Accordingly, in the high load region where the voltage control ratecommand ks** is larger than the predetermined value, the correctioncoefficient α2 is adopted as the correction coefficient α, and in thelow load region where the voltage control rate command ks** is smallerthan the predetermined value, the correction coefficient α1 is adoptedas the correction coefficient α.

<Description of Effect>

When the correction coefficient α2 which is inversely proportional tothe amplitude of the alternating current is adopted regardless of thelow load region and the high load region, the correction coefficient α2becomes extremely large in the low load region (refer to FIG. 14).Herein, for simplification, the current is considered to be small in thelow load region. In this case, the ratio of the correction coefficientto the voltage control rate command ks** increases, and the VL controlsystem may be unstable. Thus, different from the present embodiment, itis considered that the correction coefficient α is set to zero in thelow load region. In this case, the voltage control rate is not correctedin the low load region. Thus, the harmonic component of the DC voltageVdc cannot be reduced in the low load region. FIG. 15 schematicallyshows exemplary input current and DC voltage Vdc of the rectifier 1 inthe low load region when the voltage control rate is not corrected inthe low load region.

In the meanwhile, in the present embodiment, the voltage control rate ksis corrected also in the low load region. Thus, the harmonic componentof the DC voltage Vdc can be reduced also in the low load region. FIG.16 schematically shows exemplary input current and DC voltage Vdc of therectifier 1 in the low load region when the voltage control rate iscorrected in the low load region as in the present embodiment.

As can be understood from a comparison of FIGS. 15 and 16, the presentembodiment enables the reduction of the harmonic component of the DCvoltage Vdc, thereby also enabling the reduction of the harmoniccomponent of the input current.

When the correction coefficient α is set to zero in the low load region,the input current and the DC voltage Vdc change comparatively largely ata boundary between the low load region and the high load region. Thereason is that the harmonic component is large in the low load region,whereas the harmonic component decreases in the high load region. Such achange causes a vibration.

In the meanwhile, the present embodiment enables the reduction of theharmonic component in both the low load region and the high load region.Thus, a waveform of the input current and the direct current does notlargely change around the boundary between the low load region and thehigh load region. Accordingly, the above vibration can be suppressed.

The Third Embodiment

As can be understood from the equation (3), the ratio of the correctionamount H to the voltage control rate command ks** is made larger as thevoltage VL becomes larger. When this ratio is too large, the stabilityof the control may be impaired.

Thus, in the third embodiment, it is considered that a limitation is seton the correction amount H (=α·K·VL). Considering that the harmoniccomponent of the DC voltage Vdc appears in the voltage VL, the voltageVL takes positive and negative values. Thus, when the correction amountH is larger than a positive upper limit value HPlimit, the correctionamount H is limited to the upper limit value HPlimit, and whencorrection amount H is smaller than a negative lower limit valueHMlimit, the correction amount H is limited to the lower limit valueHMlimit. For example, absolute values of the upper limit value HPlimitand lower limit value HMlimit can be equal to each other. Herein, inaccordance with the above limitation, the absolute value of thecorrection amount H is limited to a limit value Hlimit(=HPlimit=|HMlimit|).

However, even when the correction amount H is limited to a certainvalue, the ratio increases as the voltage control rate command ks**becomes smaller. Thus, in the third embodiment, the limit value Hlimitis set smaller as the voltage control rate command ks** becomes smaller.For example, as shown in FIG. 18, the limit value Hlimit is set to beproportional to the voltage control rate command ks**. When this isformulated, the following equation is derived.Hlimit=B·ks**  (12)

A proportionality coefficient B is appropriately set and, for example, avalue around 0.2 to 0.25 may be adopted. Accordingly, the ratio (H/ks**)can be limited to the proportionality coefficient B or less. Thus, theinstability of the control can be suppressed.

<Configuration>

The power conversion apparatus according to the third embodiment isdifferent from the power conversion apparatus of FIG. 1 in the harmonicsuppression controller 31. FIG. 17 is a view showing an exemplaryconceptual configuration of the harmonic suppression controller 31. Theharmonic suppression controller 31 further includes a variable limiter316 compared with the harmonic suppression controller 31 of FIG. 13. Thevariable limiter 316 receives the voltage control rate command ks** fromoutside. The variable limiter 316 calculates the limit value Hlimitwhich is made larger as the voltage control rate command ks** becomeslarger. For example, the voltage control rate command ks** is multipliedby the proportionality coefficient B to calculate the limit valueHlimit.

The variable limiter 316 receives the multiplication result (α·K·VL)from the multiplier 313. When the multiplication result is larger thanthe upper limit value (the limit value Hlimit), the variable limiter 316outputs the upper limit value (the limit value Hlimit) as the correctionamount H, and when the multiplication result is smaller than the lowerlimit value (a value obtained by multiplying the limit value Hlimit by−1), the variable limiter 316 outputs the lower limit value as thecorrection amount H. When the calculation result is smaller than theupper limit value and larger than the lower limit value, the variablelimiter 316 outputs the calculation result as the correction amount H.This correction amount H is outputted to the voltage control ratecorrector 32. Accordingly, the absolute value of the correction amount Hcan be limited to the limit value Hlimit.

The above various embodiments can be appropriately combined as long asthey do not impair each other's functions.

While the disclosure has been shown and described in detail, theforegoing description is in all aspects illustrative and notrestrictive. It is therefore understood that numerous modifications andvariations can be devised without departing from the scope of thedisclosure.

The invention claimed is:
 1. A method of controlling a power conversionapparatus which comprises: a first power supply line and a second powersupply line; a rectifier for rectifying a first AC voltage to beinputted into a DC voltage to output said DC voltage to between saidfirst power supply line and said second power supply line; a capacitorprovided between said first power supply line and said second powersupply line; a reactor constituting an LC filter, together with saidcapacitor; and a power converter for converting a DC voltage supportedby said capacitor into a second AC voltage based on a switching signalto be inputted and then applying said second AC voltage to a load toflow an alternating current, said method comprising: detecting a voltageof said reactor taking a potential on a side of said capacitor as areference; when at least a first voltage control rate command which is aratio of an amplitude of said second AC voltage to an average value ofsaid DC voltage is smaller than a predetermined value, calculating acorrection coefficient which is made smaller as said first voltagecontrol rate command becomes smaller; making a correction on said firstvoltage control rate command to subtract a correction amount obtained bymultiplying said voltage of said reactor by said correction coefficient,thereby generating a second voltage control rate command; and givingsaid switching signal which is generated based on said second voltagecontrol rate command to said power converter.
 2. The method ofcontrolling the power conversion apparatus according to claim 1, whereinwhen said first voltage control rate command is larger than saidpredetermined value, said correction coefficient is calculated to bemade larger as said amplitude of said alternating current becomessmaller.
 3. The method of controlling the power conversion apparatusaccording to claim 2, wherein when said first voltage control ratecommand is larger than said predetermined value, said correctioncoefficient is inversely proportional to said amplitude of saidalternating current.
 4. The method of controlling the power conversionapparatus according to claim 1, wherein when said correction amount islarger than a positive upper limit value, said correction amount islimited to said positive upper limit value, and when said correctionamount is smaller than a negative lower limit value, said correctionamount is limited to said negative lower limit value, and absolutevalues of said positive upper limit value and said negative lower limitvalue are made larger as said first voltage control rate command becomeslarger.
 5. The method of controlling the power conversion apparatusaccording to claim 4, wherein said positive upper limit value or saidnegative lower limit value is proportional to said first voltage controlrate command.
 6. The method of controlling the power conversionapparatus according to claim 2, wherein when said correction amount islarger than a positive upper limit value, said correction amount islimited to said positive upper limit value, and when said correctionamount is smaller than a negative lower limit value, said correctionamount is limited to said negative lower limit value, and absolutevalues of said positive upper limit value and said negative lower limitvalue are made larger as said first voltage control rate command becomeslarger.
 7. The method of controlling the power conversion apparatusaccording to claim 6, wherein said positive upper limit value or saidnegative lower limit value is proportional to said first voltage controlrate command.
 8. The method of controlling the power conversionapparatus according to claim 3, wherein when said correction amount islarger than a positive upper limit value, said correction amount islimited to said positive upper limit value, and when said correctionamount is smaller than a negative lower limit value, said correctionamount is limited to said negative lower limit value, and absolutevalues of said positive upper limit value and lower said negative limitvalue are made larger as said first voltage control rate command becomeslarger.
 9. The method of controlling the power conversion apparatusaccording to claim 8, wherein said positive upper limit value or saidnegative lower limit value is proportional to said first voltage controlrate command.